The disclosed embodiments generally relate to a phase shifter design, and in particular, to a passive phase shifter with a wide bandwidth that can be used in both radio frequency and microwave applications.
In this technology age of more reliable and faster data speeds, the accompanying hardware needs to accommodate the higher data rates with higher bandwidth performance. One such component of the hardware is the phase shifter. Phase shifters have multiple applications, for example, in linearization, amplification, power mixing, power dividing, power coupling, metrology, and instrumentation, and in particular, in phase modulation communication systems and phased array antenna systems.
A classic phase shifter is the Schiffman phase shifter, as described in B. Schiffman, “A new class of broadband microwave 90-degree phase shifters,” IRE Transactions on Microwave Theory & Techniques, vol. MTT-6, no. 4, pp. 232-237, April 1958, which employs sections of coupled-strip transmission lines to create phase shift elements. Even with improvements made to the Schiffman phase shifter designs, such as those described in J. L. R. Quirarte and J. P. Starski, “Novel Schiffman phase shifters,” IEEE Transactions on Microwave Theory & Techniques, vol. 41, no. 1, pp. 9-14, January 1993, and those described in Y. Guo, Z. Zhang, and L. Ong, “Improved wideband Schiffman phase shifter,” IEEE Transactions on Microwave Theory & Techniques, vol. 54, no. 3, pp. 1196-1200, March 2006, the implementations hinge on extremely tight coupling, narrow microstrip lines, and very narrow coupling gaps, for wide bandwidth performance. For operation around 13 GHz, the dimensions of the narrow microstrip lines and coupling gaps disclosed in the previous publications require specific fabrication techniques that are not realizable with more prevalent and less expensive printed circuit board fabrication technology.
Other techniques, such as those described in A. M. Abbosh, “Ultra-wideband phase shifters,” IEEE IEEE Transactions on Microwave Theory & Techniques, vol. 55, no. 9, pp. 1935-1941, September 2007, M. Naser-Moghadasi, G. R. Dadashzadeh, A. Dadgarpour, F. Jolani, and B. S. Virdee, “Compact ultra wideband phase shifter,” Progress In Electromagnetics Research Letters, vol. 15, pp. 89-98, 2010, and M. A. Honarvar, F. Jolani, A. Dadgarpour, and B. S. Virdee, “Compact wideband phase shifter,” International Journal of RF and Microwave Computer-Aided Engineering, vol. 23, no. 1, pp. 47-51, January 2013, exploit broadside coupling among layers of microstrip patches and slots. However, these techniques add a layer of complexity with fabrication of additional layers and are not practical when integrating the phase shifter with other microwave components.
A broadband approach using a loaded transmission line concept is proposed in S. Y. Zheng, W. S. Chan, and K. F. Man, “Broadband phase shifter using loaded transmission line,” IEEE Microwave and Wireless Components Letters, vol. 20, no. 9, pp. 498-500, September 2010. As shown in FIG. 1, a half wavelength transmission line 105 was loaded in the middle with a half wavelength open stub 110. For additional bandwidth and compactness, a T-shaped stepped impedance half wavelength stub 205 was designed as shown in FIG. 2. The T shaped half wavelength stub 205 is implemented as a quarter wavelength stub terminated by quarter wavelength triangular patches. The phase characteristics are mainly determined by the length of the open stub and the triangular patches. Microstrip widths w1 and w2, and length l2 are chosen for impedance matching. The phase characteristics of both the half wavelength open stub 110 and the T shaped half wavelength stub 205 with the same characteristic impedance are shown in FIG. 3. However, the sharply acute angles of this design are difficult to manufacture, this approach is not easy to replicate for other operating frequencies, and the amount of phase ripple leaves room for improvement.
FIG. 4 shows a design as described in S. H. Yeung, Q. Xue, and K. F. Man, “Broadband 90° differential phase shifter constructed using a pair of multisection radial line stubs,” IEEE Transactions on Microwave Theory & Techniques, vol. 60, no. 9, pp. 2760-2767, September 2012. The implementation illustrated in FIG. 4 has a pair of multi-sectional radial open stubs 405, 410 that provide a 10% improvement in bandwidth and 1.4 degree improvement in phase error over the design shown in FIG. 2. However, this 10% improvement in bandwidth and 1.4 degrees of phase error costs four times more in real estate than the solution shown in FIG. 2, and while this design achieves an improvement over the design shown in FIG. 2 in terms of bandwidth and phase ripple, it is overly complex with multiple radial stublike sections.
Phase shifter designs may be realized using microstrip implementations, that is, by forming conductor shapes on a substrate. Typical conductor materials may include elemental metals such as aluminum, copper, gold, and silver, while typical substrate materials may include alumina, gallium arsenide, glass reinforced epoxy laminate, and quartz. The impedance of a microstrip line Z0 may be determined from its width and thickness, and the characteristics of the substrate. An exemplary equation is presented below for calculating the impedance of a microstrip line Z0 when the ratio of microstrip width, shown in this example as W, to substrate thickness, shown in this example as H, is less than 1. Another exemplary equation is shown when the ratio of microstrip width W, to substrate thickness H, is greater than or equal to 1:
            when      ⁢                          ⁢              (                  W          H                )              <    1              ɛ      e        =                                        ɛ            r                    +          1                2            +                                                  ɛ              r                        -            1                    2                ⁡                  [                                                    (                                  1                  +                                      12                    ⁢                                          (                                              H                        W                                            )                                                                      )                                                              -                  1                                /                2                                      +                          0.04              ⁢                                                (                                      1                    -                                          (                                              W                        H                                            )                                                        )                                2                                              ]                                Z      0        =                  60                              ɛ            eff                              ⁢              ln        ⁡                  (                                    8              ⁢                              H                W                                      +                          0.25              ⁢                              W                H                                              )                    ⁢              (        ohms        )                        when      ⁢                          ⁢              (                  W          H                )              ≥    1              ɛ      e        =                                        ɛ            r                    +          1                2            +                                                  ɛ              r                        -            1                    2                ⁢                              (                          1              +                              12                ⁢                                  (                                      H                    W                                    )                                                      )                                              -              1                        /            2                                          Z      0        =                  120        ⁢                                  ⁢        π                                          ɛ            eff                          ×                  [                                    W              H                        +            1.393            +                                          2                3                            ⁢                              ln                ⁡                                  (                                                            W                      H                                        +                    1.444                                    )                                                              ]                                    where εr is defined as the relative dielectric constant of the substrate;        εe is the effective dielectric constant of the substrate;        H is the height of the substrate; and        W equals the width of the microstrip, where it is assumed that the thickness of the microstrip t is thin enough to be ignored.        
For designs based on achieving a particular impedance, the width of the microstrip may be determined from the impedance according to the following equation:
  w  =                    7.48        ×        h                    e                  (                                    Z              0                        ⁢                                                                                ɛ                    r                                    +                  1.41                                            87                                )                      -          1.25      ×      t                      where εr is defined as the relative dielectric constant of the substrate;        h is the height of the substrate;        w equals the width of the microstrip; and        t is the thickness of the microstrip.        